High-performance mixers are essential components in radio frequency application commonly found in modern telecommunication devices.
In particular, using a direct up-conversion mixer is a highly popular approach for mixer design because of the advantages in achieving high circuit integration, small integrated circuitry area and low power consumption.
Given the advantages of the direct up-conversion mixer, the next challenge in improving the performance of this mixer is to minimize its undesirable characteristics of local oscillator leakage, local oscillator self-modulation and input-to-output signal linearity.
During the operation of a direct up-conversion mixer, a local oscillator generates a clocking signal having a frequency that is exactly equivalent to a carrier frequency for modulating a base band signal fed through a transconductance stage of the mixer. This clocking signal may however “leak” to the output of the mixer and be amplified by a power amplifier, which is typically connected to the mixer output. This results in local oscillator leakage, which corrupts the frequency spectrum of the modulated output signal of the mixer and therefore creates problems for a receiver to properly demodulate the signal.
The presence of the power amplifier together with a parasitic path within the mixer can lead to electromagnetic coupling of the modulated output signal from the power amplifier's output to the local oscillator. Such undesirable electromagnetic coupling affects the working consistency of the local oscillator and is especially prominent when higher frequencies are used. This causes the problem of local oscillator self-modulation.
The use of a band pass filter at the output of the power amplifier is not able to suppress the leaked clocking signal since this leaked clocking signal has a frequency which is exactly the same as the carrier frequency.
U.S. Pat. No. 6,370,372 to Molnar proposes a sub-harmonic mixing stage 10, referring to FIG. 1, as an exemplary way of resolving the local oscillator leakage and self-modulation problems. Molnar discloses the use of two stacked Gilbert cells 12 and 14 when designing the sub-harmonic mixing stage of a radio frequency mixer. The advantage of the sub-harmonic mixing stage is an ability to operate at one half the clocking signal frequency. The mixer can then operate on one half the carrier frequency and any local oscillator leakage may be sufficiently filtered out by a band pass filter connected to the output of the power amplifier. By filtering out any local oscillator leakage the local oscillator leakage problem is disposed off. The self-modulation problem is resolved as the frequency of the modulated output signal is significantly lower from the local oscillator frequency so that the working consistency of the local oscillator is essentially maintained.
While Molnar proposes a sub-harmonic mixing stage 10 that may be used to eliminate problems associated with local oscillator leakage and self-modulation, there are problems attendant on this proposal. Due to the circuit configuration of the sub-harmonic stage 10, which requires a two-stack topology for a three-port multiplication function, substantial voltage headroom is needed for the sub-harmonic mixing stage 10 in order for the stage to be operational. This means that for a mixer that is operating on low voltage supply, very low voltage headroom is allocated for the operation of the transconductance stage.
A high-performance mixer has to maintain output linearity under the input of a large voltage signal generated from a base band signal. The inability to meet this linearity requirement causes the output signal spectrum to exceed a particular standard spectrum mask during the mixer operation. This inability to meet the linearity requirement of the IEEE 802.11a standard can lead to interference between signals of adjacent frequencies generated by other signal sources.
Typically, in the design of the mixing stage of the direct up-conversion mixer, the sub-harmonic mixing stage is used together with a transconductance stage. The sub-harmonic mixing stage operates based on a current commutative function that is driven by a large local oscillator signal. As long as the local oscillator amplitude is sufficiently large enough, the output signal of the mixer is able to achieve linearity with the input signal from the transconductance stage. Hence, the linearity performance of the direct up-conversion mixer is highly dependent on the transconductance stage. It is therefore desirable for a direct up-conversion mixer to have a transconductance stage with high linearity and current efficiency operating under a large input range and small voltage headroom.
One technique of providing linearized transconductance is to employ a Multi-hyperbolic Tangential transconductance circuit 20, as shown in FIG. 2. This technique uses a number of asymmetric differential transistor pairs 22 and 24 with specific transistor ratios and tail currents to extend the linear range of the transconductance stage of the mixer up to seven times of thermal voltage differentially. Although the Multi-hyperbolic Tangential technique is a good approach for linearizing a small input signal, this approach has a considerably limited signal handling capability of approximately 100 mV. This undesirably distorts any base band signal with large input amplitude.
One proposed method of increasing the signal handling capability is to employ an Emitter Degeneration transconductance circuit 30, as shown in FIG. 3a. This circuit 30 enables larger signal input, with amplitude as high as one half supply voltage differentially. However, the voltage headroom needed for the circuit 30 is large, consuming about 2V DC supply. For a low supply voltage such as 2.7V, there is only 0.7V of voltage headroom remaining for the sub-harmonic stage, which is typically insufficient for such a stage to function operationally. Moreover, the current efficiency of the Emitter Degenerated transconductance circuit 30 is rather low. The current efficiency is defined as the ratio of maximum output current amplitude to the DC bias current.
With reference to FIG. 3b and FIG. 3c, the DC bias current of the Emitter Degenerated transconductance circuit 30 is 1.1 mA and the maximum output current amplitude is 0.62 mA for a third harmonic of 55 dBc under a 1.25V peak 10 MHz differential input signal. Hence, the achievable current efficiency is 55% for the Emitter Degeneration transconductance circuit 30.
Accordingly there is a need for an input transconductance stage with high input-to-output signal linearity and high current efficiency that operates under a large input signal range and small voltage headroom for a direct up-conversion mixer.